Half-bridge llc resonant converter with self-driven synchronous rectifiers

ABSTRACT

The present invention discloses a half-bridge LLC resonant converter with self-driven synchronous rectifiers, which utilizes a primary IC controller and a gate driver to drive the secondary synchronous rectifiers. In correspondence with the gate drive output voltages of the primary IC controller to the primary switch transistors, the gate driver for the secondary synchronous rectifiers comprises a differential transformer if the primary IC controller outputs two ground-referenced gate drive voltages, which cannot directly drive the primary switch transistors but can be imposed on the differential transformer; or comprises a DC shifter, a DC restorer and a differential transformer if the primary IC controller outputs two gate-source voltages, which can directly drive the primary switch transistors but cannot be imposed on the differential transformer. The drive voltages of the primary switch transistors are unipolar; however the drive voltage of the secondary synchronous rectifiers can be bipolar or unipolar. Under the valid operation mode, this converter can decrease the rectifier conduction losses to increase the power converter efficiency.

BACKGROUND OF THE INVENTION

1. Field of Invention

The present invention relates to a half-bridge LLC resonant converter with self-driven synchronous rectifiers.

2. Related Art

FIG. 1 shows a circuit diagram of a prior art. An ideal transformer T₀ includes a primary winding N_(p) and two secondary windings N_(s). A primary circuit is connected to the N_(p) and a secondary circuit to the two N_(s).

The primary circuit includes a first switch transistor M₁, a second switch transistor M₂ and an LLC resonant tank, which includes a magnetizing inductor L_(m), a resonant inductor L_(r) and a resonant capacitor C_(r). M₁ and M₂ are connected between an input voltage source V_(in) and a primary ground terminal in a half-bridge configuration, where the point at which M₁, M₂ and the LLC resonant tank intersect is called a first node P, and the LLC resonant tank is connected between the first node P and the primary ground terminal.

It is emphatically noted that a practical transformer T₁ is equivalent to the integration of the ideal transformer T₀ including the N_(p) and the two N_(s), L_(m) and a leakage inductor, where L_(m) is in parallel with the N_(p), and the leakage inductor is in series with the parallel circuit of L_(m) and N_(p). L_(m) can be measured from the primary side with the two N_(s) open-circuited, and the leakage inductance can be measured from the primary side with the two N_(s) short-circuited. If the N_(p) and the two N_(s) of T₁ are wound with a sandwich structure, then an external L_(r) is necessary, but if the N_(p) and the two N_(s) of T₁ are wound on a slotted bobbin, then the L_(r) can be provided by the leakage inductance of T₁. A transformer with a slotted bobbin is used in this example.

The secondary circuit includes a first rectifier diode D₁, a second rectifier diode D₂ and an output capacitor C_(o). D₁ and D₂ are connected in a center-tapped common-cathode rectifier configuration between the two N_(s) and C_(o), where the two N_(s) are connected in a center-tapped configuration at the secondary ground terminal, and D₁ and D₂ are connected in a common-cathode rectifier configuration at the output voltage terminal with an output voltage V_(o).

For the convenience of illustration, the circuit parameters are defined as follows: f_(s) is the switching frequency of M₁ and M₂,

$f_{r} = \frac{1}{2\pi \sqrt{L_{r}C_{r}}}$

is the resonant frequency of L_(r) and C_(r),

$n = \frac{N_{p}}{N_{s}}$

is the primary-to-secondary turns ratio of T₀, V_(o) is the output voltage, and V_(or)=nV_(o) is the reflected output voltage. Regarding the circuit variables, the reference polarities of the gate-source voltages v_(GS) ^(M1)(t) and v_(GS) ^(M2)(t) of M₁ and M₂, the resonant capacitor voltage v_(C) _(r) (t), the primary voltage v_(p)(t) and the secondary voltage v_(s)(t) as well as the reference directions of the resonant inductor current i_(L) _(r) (t), the magnetizing inductor current i_(L) _(m) (t), the primary current i_(p)(t) and the secondary current i_(s)(t) are also shown in FIG. 1.

According to the conditions of f_(s)<f_(r), f_(s)=f_(r) and f_(s)>f_(r), the waveforms of v_(GS) ^(M1)(t), v_(GS) ^(M2)(t), i_(L) _(r) (t), i_(L) _(m) (t) and i_(s)(t) are shown in FIGS. 2 a, 2 b and 2 c respectively. As shown in the figures, the waveforms of the first half period and the second half period are symmetrical, so only equivalent circuits and critical waveforms of the first half period are described and those of the second half period can be analogized using the symmetry.

Firstly, the physical meanings of t=t₀, t=t₁, t=t_(r) and t=t_(s) are interpreted as follows: t=t₀ is the time when a resonant period resumes, t=t₁ is the time when i_(L) _(r) (t) crosses 0, t=t_(r) is the time when i_(s)(t) descends to 0 and t=t_(s) is the time when v_(GS) ^(M1)(t) switches to 0.

Regardless of f_(s)≦f_(r) or f_(s)>f_(r), during the interval of t₀≦t≦t₁, v_(GS) ^(M1)(t)=0, v_(GS) ^(M2)(t)=0, i_(L) _(r) (t)<0 and i_(L) _(r) >i_(L) _(m) (t). M₁ and M₂ are turned off, i_(L) _(r) (t) flows through the body diode of M₁, i_(p)(t)=i_(L) _(r) (t)−i_(L) _(m) (t)>0 flows into the dotted terminal of N_(p), i_(s)(t)=ni_(p)>0 flows out of the dotted terminal of N_(s), D₁ is turned on and D₂ is turned off. L_(m) does not participate in the resonance of L_(r) and C_(r) due to the clamp of V_(or), i_(L) _(r) (t) and i_(s)(t) are quasi-sinusoidal waves and the rising slope of i_(L) _(m) (t) is equal to

$\frac{V_{or}}{L_{m}}.$

D₁ switches to on state at t=t₀ under zero-current-switching (ZCS), and M₁ switches to on state during t₀≦t≦t₁ under zero-voltage-switching (ZVS) and, more particularly, at t=t₁ under ZVS and ZCS, so the switching losses are reduced.

Under the condition of f_(s)≦f_(r) (i.e. t_(r)≦t_(s)), i_(s)(t) descends to 0 before M₁ turns off. The interval of t₁≦t≦t_(s) is divided into two subintervals t₁≦t≦t_(r) and t_(r)≦t≦t_(s). During the subinterval of t₁≦t≦t_(r), v_(GS) ^(M1)(t)=V_(cc), V_(GS) ^(M2)(t)=0, i_(L) _(r) (t)>0 and i_(L) _(r) >i_(L) _(m) (t). M₁ is turned on, M₂ is turned off, i_(L) _(r) (t) flows through the channel of M₁, i_(p)(t)>0 flows into the dotted terminal of N_(p), i_(s)(t)>0 flows out of the dotted terminal of N_(s), D₁ is turned on and D₂ is turned off. L_(m) does not participate in the resonance of L_(r) and C_(r) due to the clamp of V_(or), i_(L) _(r) (t) and i_(s)(t) are quasi-sinusoidal waves and the rising slope of i_(L) _(m) (t) is equal to

$\frac{V_{or}}{L_{m}}.$

D₁ switches to off state at t=t_(r) under ZCS. During the subinterval of t_(r)≦t≦t_(s), v_(GS) ^(M1)(t)=V_(cc), v_(GS) ^(M2)(t)=0, i_(L) _(r) (t)>0 and i_(L) _(r) =i_(L) _(m) (t). M₁ is turned on, M₂ is turned off, i_(L) _(r) (t) flows through the channel of M₁, i_(p)(t)=0, i_(s)(t)=0, both D₁ and D₂ are turned off. L_(m) participates the resonance of L_(r) and C_(r), and the rising slope of i_(L) _(r) (t) and i_(L) _(m) (t) is smaller than

$\frac{V_{or}}{L_{m}}.$

Especially,

${\left. {\frac{{i_{L_{m}}(t)}}{t} < \frac{V_{or}}{L_{m}}}\Rightarrow{v_{s}(t)} \right. = {{\frac{L_{m}}{n}\frac{{i_{L_{m}}(t)}}{t}} < V_{o}}},$

D₁ is reverse-biased by the voltage difference between V_(o) and v_(s)(t) to turn off, and D₂ switches to on state at t=t_(s) under ZCS.

Under the condition of f_(s)>f_(r) (i.e. t_(r)>t_(s)), i_(s)(t) descends to 0 after M₁ turns off. The interval t₁≦t≦t_(r) is divided into t₁≦t≦t_(s) and t_(s)≦t≦t_(r). During t₁≦t≦t_(s), v_(GS) ^(M1)(t)=V_(cc), v_(GS) ^(M2)(t)=0, i_(L) _(r) (t)>0 and i_(L) _(r) (t)>i_(L) _(m) (t). M₁ is turned on, M₂ is turned off, i_(L) _(r) (t) flows through the channel of M₁, i_(p)(t) flows into the dotted terminal of N_(p), i_(s)(t) flows out of the dotted terminal of N_(s), D₁ is turned on and D₂ is turned off. L_(m) does not participate in the resonance of L_(r) and C_(r) due to the clamp of V_(or), i_(L) _(r) (t) and i_(s)(t) are quasi-sinusoidal waves and the rising slope of i_(L) _(m) (t) is equal to

$\frac{V_{or}}{L_{m}}.$

During t_(s)≦t≦t_(r), v_(GS) ^(M1)(t)=0, v_(GS) ^(M2)(t)=0, i_(L) _(r) (t)>0 and i_(L) _(r) >i_(L) _(m) (t). Both M₁ and M₂ are turned off, i_(L) _(r) (t) flows through the body diode of M₂, i_(p)(t) flows into the dotted terminal of N_(p), i_(s)(t) flows out of the dotted terminal of N_(s), D₁ is truned on and D₂ is turned off. L_(m) does not participate in the resonance of L_(r) and C_(r) due to the clamp of V_(or), i_(L) _(r) (t) and i_(s)(t) are quasi-sinusoidal waves and the rising slope of i_(L) _(m) (t) is equal to V_(or)/L_(m). Especially,

${\frac{{i_{L_{m}}(t)}}{t} = {\left. \frac{V_{or}}{L_{m}}\Rightarrow{v_{s}(t)} \right. = {{\frac{L_{m}}{n}\frac{{i_{L_{m}}(t)}}{t}} = V_{o}}}},$

D₁ remains on. i_(s)(t=t_(r)) commutates from D₁ to D₂ at t=t_(r) under ZCS.

This conventional converter benefits from lower switching losses due to ZVS and ZCS but suffers from higher conduction losses due to the diode rectifiers. To reduce the conduction losses, self-driven synchronous rectifiers (SRs) are proposed. The primary switch transistors and the secondary SRs are driven by an IC controller and a gate driver simultaneously. Conceptually, the IC controller can be a primary IC controller or a secondary IC controller. Practically, a primary IC controller has three advantages over a secondary IC controller: (1) easier to buy from the market, (2) easier to cooperate with a primary power factor corrector and (3) easier to realize the protection functions of the converter. Based on a primary IC controller, a cost-effective half-bridge LLC resonant converter with self-driven synchronous rectifiers is proposed.

SUMMARY OF THE INVENTION

According to an aspect of the present invention, a half-bridge LLC resonant converter with self-driven synchronous rectifiers uses a primary IC controller and a gate driver to drive the primary switch transistors and the secondary synchronous rectifiers.

In correspondence with the gate drive output voltages of the primary IC controller to the primary switch transistors, the gate driver of the secondary synchronous rectifiers comprises a differential transformer if the primary IC controller outputs two ground-referenced gate drive voltages, which should be followed by a drive module for driving the primary switch transistors; or further comprises a DC shifter and a DC restorer if the primary IC controller outputs two drive voltages referred to the sources of two primary switch transistors and these two voltages can directly drive the primary switch transistors.

The drive voltage of the primary switch transistors is unipolar, and the drive voltage of the secondary synchronous rectifiers can be bipolar or unipolar.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a circuit diagram of a half-bridge LLC resonant converter according to a prior art.

FIGS. 2 a, 2 b and 2 c show voltage and current waveforms for the conditions of f_(s)<f_(r), f_(s)=f_(r) and f_(s)>f_(r) respectively.

FIGS. 3 a and 3 b show a circuit diagram and drive voltage waveforms of the first embodiment according to the present invention.

FIGS. 4 a and 4 c show a circuit diagram and drive voltage waveforms of the second embodiment according to the present invention.

FIGS. 4 b and 4 c show a circuit diagram and drive voltage waveforms of the third embodiment according to the present invention.

FIGS. 5 a and 5 b show a circuit diagram and drive voltage waveforms of the fourth embodiment according to the present invention.

FIGS. 6 a and 6 c show a circuit diagram and drive voltage waveforms of the fifth embodiment according to the present invention.

FIGS. 6 b and 6 c show a circuit diagram and drive voltage waveforms of the sixth embodiment according to the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Firstly, the effect of the relationship between the switching frequency and the resonant frequency on the converter operation is illustrated with the analyses of FIG. 2 a through 2 c. Six exemplary circuit topologies according to the present invention are shown in FIG. 3 a, FIG. 4 a, FIG. 4 b, FIG. 5 a, FIG. 6 a and FIG. 6 b.

Under the condition of f_(s)<f_(r), during the interval of t_(r)≦t≦t_(s), the first switch transistor M₁ is turned on and the second switch transistor M₂ is turned off, so the first synchronous rectifier SR₁ is turned on and the second synchronous rectifier SR₂ is turned off. A reverse voltage resulting from the voltage difference between the output voltage V_(o) and the secondary voltage v_(s)(t) is imposed on SR₁. The reverse voltage imposed on the conducting SR₁ will cause a huge shoot-through current

${i_{st}(t)} = \frac{V_{o} - {v_{s}(t)}}{R_{on}}$

to burn down SR₁, where R_(on) is the very small on-resistance of M₁.

Under the condition of f_(s)>f_(r), during the interval of t_(s)≦t≦t_(r), both M₁ and M₂ are turned off, so SR₁ and SR₂ are turned off. Even if the channel of SR₁ is cut off, i_(s)(t)>0 still can flow through the body diode of SR₁, the converter still can operate safely. Therefore, all the embodiments according to the present invention are merely applicable to the condition of f_(s)>f_(r).

The voltage waveforms shown in FIG. 3 b, FIG. 4 c, FIG. 5 b and FIG. 6 c correspond to the six embodiments shown in FIGS. 3 a, 4 a, 4 b, 5 a, 6 a and 6 b. It is emphatically noted that M₁, M₂, SR₁ and SR₂ according to the present invention can be implemented with a p-channel metal oxide semiconductor field effect transistor (PMOS), an n-channel metal oxide semiconductor field effect transistor (NMOS), a p-type junction field effect transistor (p-JFET) or an n-type junction field effect transistor (n-JFET). For the convenience of illustration, it is assumed in this text that M₁, M₂, SR₁ and SR₂ are all implemented with NMOS.

Three exemplary embodiments are shown in FIG. 3 a, FIG. 4 a and FIG. 4 b when the primary IC controller U₁ outputs two ground-referenced drive voltages v_(B)(t) and v_(A)(t). The circuit diagram and drive voltage waveforms of the first embodiment according to the present invention are shown in FIGS. 3 a and 3 b, respectively. The ideal transformer T₀ comprises a primary winding N_(p) and two secondary windings N_(s). A primary circuit is connected to the N_(p) and a secondary circuit to the two N_(s).

The primary circuit includes a first switch transistor M₁, a second switch transistor M₂ and an LLC resonant tank, which includes a magnetizing inductor L_(m), a resonant inductor L_(r) and a resonant capacitor C_(r). M₁ and M₂ are connected between an input voltage source V_(in) and a primary ground terminal in a half-bridge configuration, where the point at which M₁, M₂ and LLC resonant tank intersect is called a first node P with a voltage V_(P), and the LLC resonant tank is connected between the first node P and the primary ground terminal.

It is emphatically noted that a practical transformer T₁ is equivalent to the integration of the ideal transformer T₀ including the N_(p) and the two N_(s), L_(m) and a leakage inductor, where L_(m) is in parallel with the N_(p), and the leakage inductor is in series with the parallel circuit of L_(m) and N_(p). L_(m) can be measured from the primary side with the two N_(s) open-circuited, and the leakage inductance can be measured from the primary side with the two N_(s) short-circuited. If the N_(p) and the two N_(s) of T₁ are wound with a sandwich structure, then an external L_(r) is necessary, but if the N_(p) and the two N_(s) of T₁ are wound on a slotted bobbin, then the L_(r) can be provided by the leakage inductance of T₁. A transformer with a slotted bobbin is used in this example hereafter but it can be replaced by an ordinary transformer having a sandwich winding structure in series with an external L_(r).

When M₁ is turned on but M₂ is turned off, V_(P) is equal to V_(in), but when M₁ is turned off but M₂ is turned on, V_(P) is equal to 0. This means that the potential V_(P) is fluctuating. The output voltages v_(B)(t) and v_(A)(t) of U₁ are referred to the primary ground, so they cannot be directly used as the gate-source voltages v_(GS) ^(M) ¹ (t) and v_(GS) ^(M) ² (t) for M₁ and M₂, especially for M₁. In this case, an IC-based or a transformer-based driver module U₂ is needed to convert v_(B)(t) and v_(A)(t) referred to the primary ground into v_(GS) ^(M) ¹ (t) and v_(GS) ^(M) ² (t) referred to the sources to M₁ and M₂.

The secondary circuit includes a first synchronous rectifier SR₁, a second synchronous rectifier SR₂ and an output capacitor C_(o). SR₁ and SR₂ are connected in a center-tapped common-source rectifier configuration between the two N_(s) and the secondary ground terminal, where the two N_(s) are connected at the output voltage terminal and the common source of SR₁ and SR₂ is connected at the secondary ground terminal G.

SR₁ and SR₂ are driven by a differential transformer T₃, which has a primary winding and two secondary windings as well as a 1:1:1 primary-to-secondary turns ratio, so a primary bipolar differential voltage v_(T) ₃ (t)=v_(B)(t)−v_(A)(t) of T₃ generates two secondary bipolar gate-source voltages v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) of SR₁ and SR₂. v_(T) ₃ (t), v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) are listed in Table 1:

TABLE 1 ν_(T) ₃ (t) ν_(GS) ^(SR) ¹ (t) ν_(GS) ^(SR) ² (t)   V_(cc)   V_(cc) −V_(cc) 0 0 0 −V_(cc) −V_(cc)   V_(cc)

The corresponding voltage waveforms of v_(A)(t), v_(B)(t), v_(GS) ^(M) ¹ (t), v_(GS) ^(M) ² (t), v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) are shown in FIG. 3 b.

A circuit diagram of the second embodiment according to the present invention is shown in FIG. 4 a, where two half-wave rectifiers and two fast turn-off circuits are connected between the secondary windings of T₃ and the gates of SR₁ and SR₂, respectively. One of the two half-wave rectifiers comprises a diode D₅₂ and a resistor R₅ for SR₁, and the other a diode D₆₂ and a resistor R₆ for SR₂. One of the two fast turn-off circuits comprises a diode D₅₁ and a PNP bipolar transistor Q₅ for SR₁, and the other a diode D₆₁ and a PNP bipolar transistor Q₆ for SR₂.

v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) are provided by two voltages, which are first induced by the two secondary windings of T₃ and then processed by the half-wave rectifiers as well as the fast turn-off circuits. When V_(T) ₃ (t)=V_(cc), D₅₂, D₅₁ and Q₆ are turned on but Q₅, D₆₂ and D₆₁, are turned off, so SR₁ is turned on but SR₂ is turned off. When V_(T) ₃ (t)=0, D₅₂, D₅₁, D₆₂ and D₆₁ are turned off but Q₅ and Q₆ are turned on, so both SR₁ and SR₂ are turned off. When V_(T) ₃ (t)=−V_(cc), D₆₂, D₆₁ and Q₅ are turned on but Q₆, D₅₂ and D₅, are turned off, so SR₂ is turned on but SR₁ is turned off. V_(T) ₃ (t), v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) are listed in Table 2:

TABLE 2 ν_(T) ₃ (t) ν_(GS) ^(SR) ¹ (t) ν_(GS) ^(SR) ² (t)   V_(cc) V_(cc) 0 0 0 0 −V_(cc) 0 V_(cc)

A circuit diagram of the third embodiment according to the present invention is shown in FIG. 4 b. v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) are provided by a differential transformer T₅ and a signal distributor, which comprises a diode D₇ and a diode D₈. T₅ has a primary winding and a secondary winding as well as a 1:1 primary-to-secondary turns ratio, so a primary bipolar differential voltage v_(T) _(s) (t)=v_(B)(t)−v_(A)(t) or T₅ generates an identical secondary bipolar differential voltage. D₇ and D₈ are connected in a common-anode configuration between the secondary winding of T₅ and the gates of SR₁ and SR₂. The signal distributor is used for converting the secondary bipolar differential voltage into two unipolar drive voltages as well as distributing these two voltages to SR₁ and SR₂ respectively.

When v_(T) ₅ (t)=V_(cc), D₈ is turned on but D₇ is turned off, so SR₁ is turned on but SR₂ is turned off. When V_(T) ₅ (t)=0, both D₇ and D₈ are turned off, so both SR₁ and SR₂ are turned off. When V_(T) ₅ (t)=−V_(cc), D₇ is turned on but D₈ is turned off, so SR₂ is turned on but SR₁ is turned off. v_(T) ₅ (t), v_(GS) ^(SR) ₁ (t) and v_(GS) ^(SR) ² (t) are listed in Table 3, and the corresponding voltage waveforms of v_(A)(t), v_(B)(t), v_(GS) ^(M) ¹ (t), v_(GS) ^(M) ² (t), v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t) of the second and the third embodiments are shown in FIG. 4 c.

TABLE 3 ν_(T) ₅ (t) ν_(GS) ^(SR) ¹ (t) ν_(GS) ^(SR) ² (t)   V_(cc) V_(cc) 0 0 0 0 −V_(cc) 0 V_(cc)

Three exemplary embodiments are shown in FIG. 5 a, FIG. 6 a and FIG. 6 b, when the primary IC controller U₁ outputs two drive voltages referred to the sources of M₁ and M₂ for directly driving M₁ and M₂. However, the output drive voltage of U₁ for M₁ is referred to the source of M₁ but not the primary ground instead, so it cannot be directly used as v_(B)(t) on T₃, but the output drive voltage of U₁ for M₂ is referred to the primary ground, so it can be used as v_(A)(t) on T₃. In view of this, the combined circuit of a DC shifter and a DC restorer is used to convert the output drive voltage of U₁ for M₁ referred to the source of M₁ into v_(B) (t) referred to the primary ground. The DC shifter comprises a capacitor C₄ and a pulse transformer T₄ that has a primary winding and a secondary winding as well as a 1:1 primary-to-secondary turns ratio. The DC restorer comprises a capacitor C₃ and a diode D₃. T₃ is connected between the DC restorer and the gates of SR₁ and SR₂ to convert a primary bipolar voltage v_(T) ₃ (t)=v_(B)(t)−v_(A)(t) into two secondary bipolar voltages v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t).

The DC shifter converts the output drive voltage of U₁ for M₁ to an AC voltage, and then the DC restorer converts the AC voltage back to a DC voltage referred to the primary ground. The voltage across C₄ can be derived from the volt-seconds product equilibrium equation:

(V _(cc) −V _(C4))D=V _(C4)(1−D)

V _(C4) =DV _(cc)

where D is the duty ratio of M₁ and D≈0.5

V_(C4)=DV_(cc)≈0.5V_(cc), so V_(C) ₄ can be viewed as a constant voltage source during a switching period. The voltage across the secondary winding of T₄ can be expressed as:

${v_{w}(t)} = \left\{ \begin{matrix} {{V_{cc} - V_{C_{4}}},} & {M_{1}\mspace{14mu} {on}\mspace{14mu} {and}\mspace{14mu} D_{3}\mspace{14mu} {off}} \\ {{- V_{C_{4}}},} & {M_{1}\mspace{14mu} {off}\mspace{14mu} {and}\mspace{14mu} D_{3}\mspace{14mu} {on}} \end{matrix} \right.$

When D₃ is turned on, C₃ is recharged to V_(C) ₄ . Therefore, the voltage across C₃, V_(C) ₃ =V_(C) ₄ ≈0.5V_(cc), can be also viewed as a constant voltage source during a switching period.

The voltage difference between the node B and the primary ground terminal can be expressed as:

$\begin{matrix} {{v_{B}(t)} = {V_{C_{3}} + {v_{w}(t)}}} \\ {= {V_{C_{4}} + \left\{ \begin{matrix} {{V_{cc} - V_{C_{4}}},} & {M_{1}\mspace{14mu} {on}\mspace{14mu} {and}\mspace{14mu} D_{3}\mspace{14mu} {off}} \\ {{- V_{C_{4}}},} & {M_{1}\mspace{14mu} {off}\mspace{14mu} {and}\mspace{14mu} D_{3}\mspace{14mu} {on}} \end{matrix} \right.}} \\ {= \left\{ \begin{matrix} {V_{cc},} & {M_{1}\mspace{14mu} {on}\mspace{14mu} {and}\mspace{14mu} D_{3}\mspace{14mu} {off}} \\ {0,} & {M_{1}\mspace{14mu} {off}\mspace{14mu} {and}\mspace{14mu} D_{3}\mspace{14mu} {on}} \end{matrix} \right.} \end{matrix}$

The voltage of the node B is denoted as v_(B)(t) referred to the primary ground, so the differential voltage v_(T) ₃ (t)=v_(B)(t)−v_(A)(t) can be imposed on T₃ to generate v_(GS) ^(SR) ¹ (t) and v_(GS) ^(SR) ² (t).

The secondary circuit of the fourth embodiment shown in FIG. 5 a is the same as that of the first embodiment shown in FIG. 3 a, so they have similar voltage waveforms shown in FIGS. 3 b and 5 b. The fifth and sixth embodiments shown in FIG. 6 a and FIG. 6 b respectively have the same primary circuit as the fourth embodiment shown in FIG. 5 a as well as the same secondary circuit as the second and third embodiments shown in FIG. 4 a and FIG. 4 b, so they have similar voltage waveforms shown in FIGS. 4 c and 6 c. The operational principles of the fifth and the sixth embodiments can be inferred from the aforementioned embodiments, and will not be restated here.

While the invention has been described in terms of what are presently considered to be the most practical and preferred embodiments, it is to be understood that the invention needs not be limited to the disclosed embodiments. On the contrary, it is intended to cover various modifications and similar arrangements included within the spirit and scope of the appended claims which are to be accorded with the broadest interpretation so as to encompass all such modifications and similar structures. 

1. A half-bridge LLC resonant converter with self-driven synchronous rectifiers, comprising: a first switch transistor and a second switch transistor connected between an external voltage source and a primary ground terminal in a half-bridge configuration, wherein the connection point of said first switch transistor and said second switch transistor is called a first node; an LLC resonant tank connected between said first node and said primary ground terminal, wherein said LLC resonant tank comprises a resonant capacitor, a resonant inductor, and a magnetizing inductor, and said magnetizing inductor provided by a primary winding of a power transformer; a power loop connecting between a first secondary winding and a second secondary winding of said power transformer, wherein said power loop comprises a first synchronous rectifier transistor, a second synchronous rectifier transistor and a filter capacitor, said first synchronous rectifier transistor and said second synchronous rectifier transistor are connected in a center-tapped common-source rectifier configuration between said first and said second secondary windings of said power transformer and a secondary ground terminal, the other two terminals of said first secondary winding and said second secondary winding are connected to a voltage output terminal, and said capacitor is connected between said voltage output terminal and said secondary ground terminal; a primary IC controller; and a gate driver connected between said primary IC controller and gates of said first synchronous rectifier transistor and said second synchronous rectifier transistor.
 2. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 1, wherein said gate driver comprises a differential transformer, a DC shifter and a DC restorer, said differential transformer comprises a primary winding and two secondary windings, said DC shifter is connected to said primary IC controller, said DC restorer is connected to primary winding of said differential transformer, said two secondary windings are constructed in a center-tapped configuration at said secondary ground terminal and connected to gates of said first synchronous rectifier and said second synchronous rectifier, respectively.
 3. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 2, further comprising two combination circuits respectively connected between secondary windings of said differential transformer and gates of said first synchronous rectifier transistor and said second synchronous rectifier transistor, and each of said two combination circuits comprises a half-wave rectifier and a fast turn-off circuit.
 4. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 3, wherein said half-wave rectifier comprises a diode and a resistor and said fast turn-off circuit comprises a diode and a PNP bipolar transistor.
 5. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 2, wherein said DC shifter comprises a capacitor and a pulse transformer and said DC restorer comprises a capacitor and a diode.
 6. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 1, wherein said gate driver comprises a differential transformer, a DC shifter, a DC restorer and a signal distributor, said differential transformer comprises a primary winding and a secondary winding, said DC shifter is connected to said primary IC controller, said DC restorer is connected to primary winding of said differential transformer, two terminals of said secondary winding of said differential transformer are respectively connected to gates of said first synchronous rectifier and said second synchronous rectifier, said signal distributor is constructed in a common-anode configuration at the secondary ground terminal and connected between two terminals of said secondary winding of said differential transformer.
 7. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 6, wherein said DC shifter comprises a capacitor and a pulse transformer and said DC restorer comprises a capacitor and a diode.
 8. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 1, further comprising a driving module for converting the output voltages of said primary IC controller into the drive voltages of said first switch transistor and said second switch transistor.
 9. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 8, wherein said gate driver comprises a differential transformer, said differential transformer comprises a primary winding and two secondary windings, said primary winding is connected to the primary IC controller, said two secondary windings are constructed in center-tapped configuration at said secondary ground terminal and connected to gates of said first synchronous rectifier and said second synchronous rectifier, respectively.
 10. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 9, further comprising two combination circuits respectively connected between secondary windings of said differential transformer and gates of said first synchronous rectifier transistor and said second synchronous rectifier transistor, and each of said two combination circuits comprises a half-wave rectifier and a fast turn-off circuit.
 11. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 10, wherein said half-wave rectifier comprises a diode and a resistor and said fast turn-off circuit comprises a diode and a PNP bipolar transistor.
 12. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 8, wherein said gate driver comprises a differential transformer and a signal distributor, said differential transformer comprises a primary winding and a secondary winding, said primary winding of said differential transformer is connected to said primary IC controller, two terminals of said secondary winding of said differential transformer are respectively connected to gates of said first synchronous rectifier and said second synchronous rectifier, and said signal distributor is constructed in a common-anode configuration at the secondary ground terminal and connected between two terminals of said secondary winding of said differential transformer.
 13. The half-bridge LLC resonant converter with self-driven synchronous rectifiers according to claim 8, wherein said driving module is IC-based or transformer-based. 